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LT6109-1/lt6109-2 1 610912fa typical application features description high side current sense amplifer with reference and comparators the lt ? 6109 is a complete high side current sense device that incorporates a precision current sense amplifer, an integrated voltage reference and two comparators. two versions of the lt6109 are available. the LT6109-1 has the comparators connected in opposing polarity and the lt6109-2 has the comparators connected in the same polar - ity. in addition, the current sense amplifer and comparator inputs and outputs are directly accessible. the amplifer gain and comparator trip points are confgured by external resistors. the open-drain comparator outputs allows for easy interface to other system components. the overall propagation delay of the lt6109 is typically only 1.4s, allowing for quick reaction to overcurrent and undercurrent conditions. the 1mhz bandwidth al - lows the lt6109 to be used for error detection in critical applications such as motor control. the high threshold accuracy of the comparators, combined with the ability to latch both comparators, ensures the lt6109 can capture high speed events. the lt6109 is fully specifed for operation from C40c to 125c, making it suitable for industrial and automotive ap - plications. the lt6109 is available in a small 10-lead msop. circuit fault protection with latching load disconnect and early warning indication applications n current sense amplifer C fast step response: 500ns C low offset voltage: 125v maximum C low gain error: 0.2% maximum n internal 400mv precision reference n internal latching comparators with reset C fast response time: 500ns C total threshold error: 1.25% maximum C two comparator polarity options n wide supply range: 2.7v to 60v n supply current: 550a n low shutdown current: 5a maximum n specifed for C40c to 125c temperature range n available in 10-lead msop package n overcurrent, undercurrent and fault detection n current shunt measurement n battery monitoring n motor control n automotive monitoring and control n remote sensing n industrial control l , lt, ltc, ltm, timerblox, linear technology and the linear logo are registered trademarks of linear technology corporation. all other trademarks are the property of their respective owners. response to overcurrent event sensehi senselo outa lt6109-2 inc2 reset inc1 2n2700 100ma warning 250ma disconnect *cmh25234b v + en/rst outc1 outc2 v ? 0.1 irf9640 3.3v 6.2v* 12v 100 6.04k 100k 1.62k10k 1k 1k 0.1f v out 2.37k 1.6k 610912 ta01a to load 5s/div v load 10v/div v outc1 5v/div i load 200ma/div 0v 0v v outc2 5v/div 0v 0ma 610912 ta01b 250ma disconnect 100ma warning
LT6109-1/lt6109-2 2 610912fa pin configuration absolute maximum ratings total supply voltage (v + to v C ) ................................. 60v maximum voltage (senselo, sensehi, outa) ............................... v + + 1v maximum v + C (senselo or sensehi) .................... 33v maximum en/ rst voltage ........................................ 60v maximum comparator input voltage ........................ 60v maximum comparator output voltage ...................... 60v input current (note 2) .......................................... C10ma sensehi, senselo input current ....................... 10ma differential sensehi or senselo input current ... 2.5ma amplifer output short-circuit duration (to v C ) .. indefnite operating temperature range (note 3) lt6109i ................................................ C40c to 85c lt6109h ............................................ C40c to 125c specifed temperature range (note 3) lt6109i ................................................ C40c to 85c lt6109h ............................................ C40c to 125c maximum junction temperature .......................... 150c storage temperature range .................. C65c to 150c lead temperature (soldering, 10 sec) ................... 300c (note 1) 1 2 3 4 5 senselo en/rst outc2 outc1 v ? 10 9 8 7 6 sensehi v + outa inc2 inc1 top view ms package 10-lead plastic msop ja = 160c/w, jc = 45c/w order information lead free finish tape and reel part marking* package description specified temperature range lt6109aims-1#pbf lt6109aims-1#trpbf ltfnj 10-lead plastic msop C40c to 85c lt6109ims-1#pbf lt6109ims-1#trpbf ltfnj 10-lead plastic msop C40c to 85c lt6109ahms-1#pbf lt6109ahms-1#trpbf ltfnj 10-lead plastic msop C40c to 125c lt6109hms-1#pbf lt6109hms-1#trpbf ltfnj 10-lead plastic msop C40c to 125c lt6109aims-2#pbf lt6109aims-2#trpbf lt fwy 10-lead plastic msop C40c to 85c lt6109ims-2#pbf lt6109ims-2#trpbf lt fwy 10-lead plastic msop C40c to 85c lt6109ahms-2#pbf lt6109ahms-2#trpbf lt fwy 10-lead plastic msop C40c to 125c lt6109hms-2#pbf lt6109hms-2#trpbf lt fwy 10-lead plastic msop C40c to 125c consult ltc marketing for parts specifed with wider operating temperature ranges. *the temperature grade is identifed by a label on the shipping container. consult ltc marketing for information on non-standard lead based fnish parts. for more information on lead free part marking, go to: http://www.linear.com/leadfree/ for more information on tape and reel specifcations, go to: http://www.linear.com/tapeandreel/
LT6109-1/lt6109-2 3 610912fa electrical characteristics the l denotes the specifcations which apply over the full operating temperature range, otherwise specifcations are at t a = 25c. v + = 12v, v pullup = v + , v en/rst = 2.7v, r in = 100, r out = r1 + r2 + r3 = 10k, gain = 100, r c = 25.5k, c l = c lc = 2pf, unless otherwise noted. (see figure 3) symbol parameter conditions min typ max units v + supply voltage range l 2.7 60 v i s supply current (note 4) v + = 2.7v, r in = 1k, v sense = 5mv 475 a v + = 60v, r in = 1k, v sense = 5mv l 600 700 1000 a a supply current in shutdown v + = 2.7v, v en/rst = 0v, r in = 1k, v sense = 0.5v l 3 5 7 a a v + = 60v, v en/rst = 0v, r in = 1k, v sense = 0.5v l 7 11 13 a a en/rst pin current v en/rst = 0v, v + = 60v C200 na v ih en/rst pin input high v + = 2.7v to 60v l 1.9 v v il en/rst pin input low v + = 2.7v to 60v l 0.8 v current sense amplifer v os input offset voltage v sense = 5mv, lt6109a v sense = 5mv, lt6109 v sense = 5mv, lt6109a v sense = 5mv, lt6109 l l C125 C350 C250 C450 125 350 250 450 v v v v ?v os /?t input offset voltage drift v sense = 5mv l 0.8 v/c i b input bias current (senselo, sensehi) v + = 2.7v to 60v l 60 300 350 na na i os input offset current v + = 2.7v to 60v 5 na i outa output current (note 5) l 1 ma psrr power supply rejection ratio (note 6) v + = 2.7v to 60v l 120 114 127 db db cmrr common mode rejection ratio v + = 36v, v sense = 5mv, v icm = 2.7v to 36v 125 db v + = 60v, v sense = 5mv, v icm = 27v to 60v l 110 103 125 db db v sense(max) full-scale input sense voltage (note 5) r in = 500 l 500 mv gain error (note 7) v + = 2.7v to 12v v + = 12v to 60v, v sense = 5mv to 100mv l C0.2 C0.08 0 % % senselo voltage (note 8) v + = 2.7v, v sense = 100mv, r out = 2k v + = 60v, v sense = 100mv l l 2.5 27 v v output swing high (v + to v outa ) v + = 2.7v, v sense = 27mv l 0.2 v v + = 12v, v sense = 120mv l 0.5 v bw signal bandwidth i out = 1ma i out = 100a 1 140 mhz khz t r input step response (to 50% of final output voltage) v + = 2.7v, v sense = 24mv step, output rising edge v + = 12v to 60v, v sense = 100mv step, output rising edge 500 500 ns ns t settle settling time to 1% v sense = 10mv to 100mv, r out = 2k 2 s
LT6109-1/lt6109-2 4 610912fa electrical characteristics the l denotes the specifcations which apply over the full operating temperature range, otherwise specifcations are at t a = 25c. v + = 12v, v pullup = v + , v en/rst = 2.7v, r in = 100, r out = r1 + r2 + r3 = 10k, gain = 100, r c = 25.5k, c l = c lc = 2pf, unless otherwise noted. (see figure 3) symbol parameter conditions min typ max units reference and comparator v th(r) (note 9) rising input threshold voltage (LT6109-1 comparator 1 lt6109-2 both comparators) v + = 2.7v to 60v, lt6109a v + = 2.7v to 60v, lt6109 l l 395 392 400 400 405 408 mv mv v th(f) (note 9) falling input threshold voltage (LT6109-1 comparator 2) v + = 2.7v to 60v, lt6109a v + = 2.7v to 60v, lt6109 l l 395 392 400 400 405 408 mv mv v hys v hys = v th(r) C v th(f) v + = 2.7v to 60v 3 10 15 mv comparator input bias current v inc1,2 = 0v, v + = 60v l C50 na v ol output low voltage i outc1,c2 = 500a, v + = 2.7v l 60 150 220 mv mv high to low propagation delay 5mv overdrive 100mv overdrive 3 0.5 s s output fall time 0.08 s t reset reset time 0.5 s t rpw valid rst pulse width l 2 15 s note 1: stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. exposure to any absolute maximum rating condition for extended periods may affect device reliability and lifetime. note 2: input and output pins have esd diodes connected to ground. the sensehi and senselo pins have additional current handling capability specifed as sensehi, senselo input current. note 3: the lt6109i is guaranteed to meet specifed performance from C40c to 85c. lt6109h is guaranteed to meet specifed performance from C40c to 125c. note 4: supply current is specifed with the comparator outputs high. when the comparator outputs go low the supply current will increase by 75a typically per comparator. note 5: the full-scale input sense voltage and the maximum output current must be considered to achieve the specifed performance. note 6: supply voltage and input common mode voltage are varied while amplifer input offset voltage is monitored. note 7: specifed gain error does not include the effects of external resistors r in and r out . although gain error is only guaranteed between 12v and 60v, similar performance is expected for v + < 12v, as well. note 8: refer to senselo, sensehi range in the applications information section for more information. note 9: the input threshold voltage which causes the output voltage of the comparator to transition from high to low is specifed. the input voltage which causes the comparator output to transition from low to high is the magnitude of the difference between the specifed threshold and the hysteresis.
LT6109-1/lt6109-2 5 610912fa typical performance characteristics input offset voltage vs temperature amplifer offset voltage vs supply voltage offset voltage drift distribution amplifer gain error vs temperature amplifer gain error distribution supply current vs supply voltage start-up supply current enable/disable response performance characteristics taken at t a = 25c, v + = 12v, v pullup = v + , v en/rst = 2.7v, r in = 100, r out = r1 + r2 + r3 = 10k, gain = 100, r c = 25.5k, c l = c lc = 2pf, unless otherwise noted. (see figure 3) supply voltage (v) 0 700 600 500 400 300 200 100 0 30 50 610912 g01 10 20 40 60 supply current (a) supply voltage (v) 0 ?100 offset voltage (v) ?60 ?20 20 10 20 30 40 610912 g05 50 60 100 ?80 ?40 0 40 80 60 5 typical units 0v v + 5v/div i s 500a/div 0a 10s/div 610912 g02 100s/div v en/rst 2v/div i s 500a/div 0v 0a 610912 g03 temperature (c) ?40 input offset voltage (v) 300 200 100 0 ?100 ?200 ?300 80 610912 g04 ?10 20 50 125110 65 ?25 5 35 95 5 typical units offset voltage drift (v/c) 6 8 12 610912 g06 4 2 0?0.5 0.5 ?1 1 1.5 2 ?1.5?2 0 10 percentage of units (%) temperature (c) ?50 ?25 ?0.20 gain error (%) ?0.10 0.05 0 50 75 610912 g07 ?0.15 0 ?0.05 25 100 125 r in = 1k r in = 100 v sense = 5mv to 100mv gain error (%) ?0.048 0 percentage of units (%) 5 15 20 25 v sense = 5mv to 100mv ?0.052 ?0.056 610912 g08 10 ?0.060 ?0.68 ?0.064 amplifer output swing vs temperature temperature (c) ?50 0 v + ? v outa (v) 0.05 0.15 0.20 0.25 0.50 0.35 0 50 75 610912 g18 0.10 0.40 0.45 0.30 ?25 25 100 125 v + = 12v v sense = 120mv v + = 2.7v v sense = 27mv
LT6109-1/lt6109-2 6 610912fa performance characteristics taken at t a = 25c, v + = 12v, v pullup = v + , v en/rst = 2.7v, r in = 100, r out = r1 + r2 + r3 = 10k, gain = 100, r c = 25.5k, c l = c lc = 2pf, unless otherwise noted. (see figure 3) amplifer input bias current vs temperature amplifer step response (v sense = 0mv to 100mv) amplifer step response (v sense = 0mv to 100mv) amplifer step response (v sense = 10mv to 100mv) amplifer step response (v sense = 10mv to 100mv) amplifer gain vs frequency system step response common mode rejection ratio vs frequency typical performance characteristics frequency (hz) 1 0 common mode rejection ratio (db) 120 100 140 10 100 1k 10k 100k 1m 10m 610912 g10 80 60 40 20 frequency (hz) 22 gain (db) 28 34 40 46 1k 100k 1m 10m 610912 g11 16 10k i outa = 1ma i outa = 100a g = 100 g = 50, r out = 5k g = 20, r out = 2k 0v v sense 100mv/div v outa 1v/div v outc1 2v/div v en/rst 5v/div 0v 0v 0v 610912 g12 2s/div r out = 2k,100mv inc1 overdrive temperature (c) ?25 input bias current (na) 60 80 100 95 610912 g13 40 20 50 70 90 30 10 0 5 35 65 ?10?40 110 20 50 80 125 sensehi senselo v outa 2v/div v sense 50mv/div 0v 0v 610912 g14 2s/div r in = 100 g = 100v/v v outa 2v/div v sense 50mv/div 0v 0v 610912 g15 2s/div r in = 100 g = 100v/v 0v 0v v outa 1v/div v sense 100mv/div 610912 g16 2s/div r in = 1k r out = 20k g = 20v/v 0v 0v v outa 1v/div v sense 100mv/div 610912 g17 2s/div r in = 1k r out = 20k g = 20v/v power supply rejection ratio vs frequency frequency (hz) 1 0 power supply rejection ratio (db) 120 100 140 160 10 100 1k 10k 100k 1m 10m 610912 g09 80 60 40 20
LT6109-1/lt6109-2 7 610912fa performance characteristics taken at t a = 25c, v + = 12v, v pullup = v + , v en/rst = 2.7v, r in = 100, r out = r1 + r2 + r3 = 10k, gain = 100, r c = 25.5k, c l = c lc = 2pf, unless otherwise noted. (see figure 3) hysteresis distribution hysteresis vs temperature hysteresis vs supply voltage comparator input bias current vs input voltage comparator input bias current vs input voltage comparator threshold distribution comparator threshold vs temperature en/rst current vs voltage comparator output low voltage vs output sink current comparator input voltage (v) ?20 comparator input bias current (na) ?10 0 10 ?15 ?5 5 20 40 610912 g25 60 0 125c 25c ?40c comparator input voltage (v) ?20 comparator input bias current (na) ?10 0 10 ?15 ?5 5 0.2 0.4 0.6 0.8 610912 g26 1.0 0 125c 25c ?40c 0 0 v ol outc1, outc2 (v) 0.25 0.50 0.75 1.00 1 i outc (ma) 2 610912 g27 3 125c 25c ?40c typical performance characteristics temperature (c) ?40 ?25 comparator threshold (mv) 398 400 402 50 110 610912 g20 396 394 392 ?10 5 20 35 80 65 125 95 404 406 408 5 typical parts comparator hysteresis (mv) 3.0 0 percentage of units (%) 5 10 15 20 30 ?40c 25c 125c 4.6 6.2 7.7 9.3 10.9 12.5 14.1 610912 g21 15.7 17.3 25 v + (v) 0 14 12 10 8 6 4 2 0 30 50 610912 g23 10 20 40 60 comparator hysteresis (mv) 5 typical parts en/rst voltage (v) 0 ?250 en/rst current (na) ?200 ?150 ?100 ?50 50 10 20 30 40 610912 g24 50 60 0 comparator threshold (mv) 0 percentage of units (%) 5 15 20 25 399.2 404 610912 g19 10 396 397.6 400.8 402.8 temperature (c) ?40 comparator hysteresis (mv) 20 18 16 14 12 10 8 6 4 0 2 80 610912 g22 ?10 20 50 125110 65 ?25 5 35 95
LT6109-1/lt6109-2 8 610912fa comparator rise/fall time vs pull-up resistor comparator step response (5mv inc1 overdrive) comparator step response (100mv inc1 overdrive) comparator reset response senselo (pin 1): sense amplifer input. this pin must be tied to the load end of the sense resistor. en/rst (pin 2): enable and latch reset input. when the en/rst pin is pulled high the lt6109 is enabled. when the en/rst pin is pulled low for longer than typically 40s, the lt6109 will enter the shutdown mode. pulsing this pin low for between 2s and 15s will reset the comparators of the lt6109. comparator propagation delay vs input overdrive pin functions outc2 (pin 3): open-drain comparator 2 output. off- state voltage may be as high as 60v above v C , regardless of v + used. outc1 (pin 4): open-drain comparator 1 output. off- state voltage may be as high as 60v above v C , regardless of v + used. v C (pin 5): negative supply pin. this pin is normally con- nected to ground. comparator output leakage current vs pull-up voltage comparator output pull-up voltage (v) 0 ?2 outc1, outc2 leakage current (na) 3 8 13 18 23 125c 10 20 30 40 610912 g28 50 60 ?40c and 25c r c pull-up resistor (k) 1 10 rise/fall time (ns) 100 1000 10000 10 100 1000 610912 g30 rise time fall time v oh = 0.9 ? v pullup v ol = 0.1 ? v pullup 100mv inc1 overdrive c l = 2pf v inc 0.5v/div 0v v outc 2v/div 0v v en/rst 5v/div 0v 610912 g31 5s/div 0v v inc 0.5v/div v outc 2v/div v en/rst 5v/div 0v 0v 610912 g32 5s/div 0v v outc 5v/div v en/rst 2v/div 0v 5s/div 610912 g33 performance characteristics taken at t a = 25c, v + = 12v, v pullup = v + , v en/rst = 2.7v, r in = 100, r out = r1 + r2 + r3 = 10k, gain = 100, r c = 25.5k, c l = c lc = 2pf, unless otherwise noted. (see figure 3) typical performance characteristics comparator input overdrive (mv) 0 comparator propagation delay (s) 3.0 4.0 5.0 160 610912 g29 2.0 1.0 2.5 3.5 4.5 1.5 0.5 0 40 80 120 200 h to l l to h
LT6109-1/lt6109-2 9 610912fa pin functions inc1 (pin 6): this is the inverting input of comparator 1. the second input of this comparator is internally connected to the 400mv reference. inc2 (pin 7): this is the input of comparator 2. for the LT6109-1 this is the noninverting input of comparator 2. for the lt6109-2 this is the inverting input of compara - tor?2. the second input of each of these comparators is internally connected to the 400mv reference. outa (pin 8): current output of the sense amplifer. this pin will source a current that is equal to the sense voltage divided by the external gain setting resistor, r in . v + (pin 9): positive supply pin. the v + pin can be con - nected directly to either side of the sense resistor, r sense . when v + is tied to the load end of the sense resistor, the sensehi pin can go up to 0.2v above v + . supply current is drawn through this pin. sensehi (pin 10): sense amplifer input. the internal sense amplifer will drive sensehi to the same potential as senselo. a resistor (typically r in ) tied from supply to sensehi sets the output current, i out = v sense /r in , where v sense is the voltage developed across r sense . block diagrams figure 1. LT6109-1 block diagram (comparators with opposing polarity) 100 outa ? + ? + 9 10 1 8 inc2 7 inc1 610912 f01 6 v + v ? v ? v ? v ? v + 3k v + 3k sensehi LT6109-1 senselo 200na reset undercurrent flag overcurrent flag 2 en/rst 3 outc2 4 outc1 34v 6v enable and reset timing ? + v ? v + 5 400mv reference
LT6109-1/lt6109-2 10 610912fa block diagrams figure 2. lt6109-2 block diagram (comparators with the same polarity) applications information the lt6109 high side current sense amplifer provides accurate monitoring of currents through an external sense resistor. the input sense voltage is level-shifted from the sensed power supply to a ground referenced output and is amplifed by a user-selected gain to the output. the output voltage is directly proportional to the current fow - ing through the sense resistor. the lt6109 comparators have a threshold set with a built-in 400mv precision reference and have 10mv of hysteresis. the open-drain outputs can be easily used to level shift to digital supplies. amplifer theory of operation an internal sense amplifer loop forces sensehi to have the same potential as senselo as shown in figure 3. connecting an external resistor, r in , between sensehi and v supply forces a potential, v sense , across r in . a corresponding current, i outa , equal to v sense /r in , will fow through r in . the high impedance inputs of the sense amplifer do not load this current, so it will fow through an internal mosfet to the output pin, outa. 100 outa ? + ? + 9 10 1 8 inc2 7 inc1 610912 f02 6 v + v ? v ? v ? v ? v + 3k v + 3k sensehi lt6109-2 senselo 200na reset overcurrent flag overcurrent flag 2 en/rst 3 outc2 4 outc1 34v 6v enable and reset timing ? + v ? v + 5 400mv reference
LT6109-1/lt6109-2 11 610912fa applications information the output current can be transformed back into a voltage by adding a resistor from outa to v C (typically ground). the output voltage is then: v out = v C + i outa ? r out where r out = r1 + r2 + r3 as shown in figure 3. table 1. example gain confgurations gain r in r out v sense for v out = 5v i outa at v out = 5v 20 499 10k 250mv 500a 50 200 10k 100mv 500a 100 100 10k 50mv 500a useful equations input voltage: v sense = i sense ? r sense voltage gain: v out v sense = r out r in current gain: i outa i sense = r sense r in note that v sense(max) can be exceeded without damag - ing the amplifer, however, output accuracy will degrade as v sense exceeds v sense(max) , resulting in increased output current, i outa . selection of external current sense resistor the external sense resistor, r sense , has a signifcant effect on the function of a current sensing system and must be chosen with care. first, the power dissipation in the resistor should be considered. the measured load current will cause power dissipation as well as a voltage drop in r sense . as a result, the sense resistor should be as small as possible while still providing the input dynamic range required by the measurement. note that the input dynamic range is the difference between the maximum input signal and the minimum accurately reproduced signal, and is limited primarily by input dc offset of the internal sense ampli- fer of the lt6109. to ensure the specifed performance, r sense should be small enough that v sense does not exceed v sense(max) under peak load conditions. as an example, an application may require the maximum sense voltage be 100mv. if this application is expected to draw 2a at peak load, r sense should be set to 50m. once the maximum r sense value is determined, the mini - mum sense resistor value will be set by the resolution or dynamic range required. the minimum signal that can be accurately represented by this sense amplifer is limited by the input offset. as an example, the lt6109 has a maximum input offset of 125v. if the minimum current is 20ma, a sense resistor of 6.25m will set v sense to 125v. this is the same value as the input offset. a larger sense resistor will reduce the error due to offset by increasing the sense voltage for a given load current. choosing a 50m r sense will maximize the dynamic range and provide a system that has 100mv across the sense resistor at peak load (2a), while input offset causes an error equivalent to only 2.5ma of load current. in the previous example, the peak dissipation in r sense is 200mw. if a 5m sense resistor is employed, then the effective current error is 25ma, while the peak sense voltage is reduced to 10mv at 2a, dissipating only 20mw. the low offset and corresponding large dynamic range of the lt6109 make it more fexible than other solutions in this respect. the 125v maximum offset gives 72db of dynamic range for a sense voltage that is limited to 500mv max. sense resistor connection kelvin connection of the sensehi and senselo inputs to the sense resistor should be used in all but the lowest power applications. solder connections and pc board interconnections that carry high currents can cause sig - nifcant error in measurement due to their relatively large resistances. one 10mm 10mm square trace of 1oz copper is approximately 0.5m. a 1mv error can be caused by as little as 2a fowing through this small interconnect. this will cause a 1% error for a full-scale v sense of 100mv. a 10a load current in the same interconnect will cause a 5% error for the same 100mv signal. by isolating the sense traces from the high current paths, this error can be reduced by orders of magnitude. a sense resistor with integrated kelvin sense terminals will give the best results. figure 3 illustrates the recommended method for connect - ing the sensehi and senselo pins to the sense resistor.
LT6109-1/lt6109-2 12 610912fa applications information selection of external input gain resistor, r in r in should be chosen to allow the required speed and resolution while limiting the output current to 1ma. the maximum value for r in is 1k to maintain good loop sta - bility. for a given v sense , larger values of r in will lower power dissipation in the lt6109 due to the reduction in i out while smaller values of r in will result in faster response time due to the increase in i out . if low sense currents must be resolved accurately in a system that has a very wide dynamic range, a smaller r in may be used if the maximum i outa current is limited in another way, such as with a schottky diode across r sense (figure 4). this will reduce the high current measurement accuracy by limiting the result, while increasing the low current measurement resolution. this approach can be helpful in cases where occasional bursts of high currents can be ignored. care should be taken when designing the board layout for r in , especially for small r in values. all trace and inter - connect resistances will increase the effective r in value, causing a gain error. the power dissipated in the sense resistor can create a thermal gradient across a printed circuit board and con - sequently a gain error if r in and r out are placed such that they operate at different temperatures. if signifcant power is being dissipated in the sense resistor then care figure 3. LT6109-1 typical connection outa i outa ? + ? + v + c1 sensehi inc2 inc1 5 4 3 2 1 r1* 610912 f03 v ? v + v + v ? LT6109-1 senselo en/rst outc2 v reset r c v pullup load v supply v sense r sense undercurrent flag overcurrent flag r in + ? outc1 *r out = r1 + r2 + r3 ? + v ? v ? v + i sense = v sense r sense r c r2* 6 7 8 9 10 r3* c l v out 400mv reference c lc c lc d sense r sense v + load 610912 f04 figure 4. shunt diode limits maximum input voltage to allow better low input resolution without overranging
LT6109-1/lt6109-2 13 610912fa applications information in this case, the only error is due to external resistor mismatch, which provides an error in gain only. however, offset voltage, input bias current and fnite gain in the amplifer can cause additional errors: output voltage error, ?v out(vos) , due to the amplifer dc offset voltage, v os ? v out(vos) = v os ? r out r in the dc offset voltage of the amplifer adds directly to the value of the sense voltage, v sense . as v sense is increased, accuracy improves. this is the dominant error of the system and it limits the available dynamic range. output voltage error, ?v out(ibias) , due to the bias currents i b + and i b C the amplifer bias current i b + fows into the senselo pin while i b C fows into the sensehi pin. the error due to i b is the following: ? v out(ibias) = r out i b + ? r sense r in ?i b ? ? ? ? ? ? ? since i b + i b C = i bias , if r sense << r in then, ?v out(ibias) = Cr out (i bias ) it is useful to refer the error to the input: ?v vin(ibias) = Cr in (i bias ) for instance, if i bias is 100na and r in is 1k, the input re - ferred error is 100v. this error becomes less signifcant as the value of r in decreases. the bias current error can be reduced if an external resistor, r in + , is connected as shown in figure 5, the error is then reduced to: v out(ibias) = r out ? i os ; i os = i b + C i b C minimizing low current errors will maximize the dynamic range of the circuit. should be taken to place r in and r out such that the gain error due to the thermal gradient is minimized. selection of external output gain resistor, r out the output resistor, r out , determines how the output cur - rent is converted to voltage. v out is simply i outa ? r out . typically, r out is a combination of resistors confgured as a resistor divider which has voltage taps going to the comparator inputs to set the comparator thresholds. in choosing an output resistor, the maximum output volt - age must frst be considered. if the subsequent circuit is a buffer or adc with limited input range, then r out must be chosen so that i outa(max) ? r out is less than the allowed maximum input range of this circuit. in addition, the output impedance is determined by r out . if another circuit is being driven, then the input impedance of that circuit must be considered. if the subsequent circuit has high enough input impedance, then almost any use - ful output impedance will be acceptable. however, if the subsequent circuit has relatively low input impedance, or draws spikes of current such as an adc load, then a lower output impedance may be required to preserve the accuracy of the output. more information can be found in the output filtering section. as an example, if the input impedance of the driven circuit, r in(driven) , is 100 times r out , then the accuracy of v out will be reduced by 1% since: v out = i outa ? r out ? r in(driven) r out + r in(driven) = i outa ? r out ? 100 101 = 0.99 ? i outa ? r out amplifer error sources the current sense system uses an amplifer and resistors to apply gain and level-shift the result. consequently, the output is dependent on the characteristics of the amplifer, such as gain error and input offset, as well as the matching of the external resistors. ideally, the circuit output is: v out = v sense ? r out r in ; v sense = r sense ? i sense
LT6109-1/lt6109-2 14 610912fa applications information there is also power dissipated due to the quiescent power supply current: p s = i s ? v + the comparator output current fows into the comparator output pin and out of the v C pin. the power dissipated in the lt6109 due to each comparator is often insignifcant and can be calculated as follows: p outc1,c2 = (v outc1,c2 C v C ) ? i outc1,c2 the total power dissipated is the sum of these dissipations: p total = p outa + p outc1 + p outc2 + p s at maximum supply and maximum output currents, the total power dissipation can exceed 100mw. this will cause signifcant heating of the lt6109 die. in order to prevent damage to the lt6109, the maximum expected dissipation in each application should be calculated. this number can be multiplied by the ja value, 160c/w, to fnd the maximum expected die temperature. proper heat sinking and thermal relief should be used to ensure that the die temperature does not exceed the maximum rating. output filtering the ac output voltage, v out , is simply i outa ? z out . this makes fltering straightforward. any circuit may be used which generates the required z out to get the desired flter response. for example, a capacitor in parallel with r out will give a lowpass response. this will reduce noise at the output, and may also be useful as a charge reservoir to keep the output steady while driving a switching circuit such as a mux or adc. this output capacitor in parallel with r out will create an output pole at: f ?3db = 1 2 ? ? r out ? c l senselo, sensehi range the difference between v batt (see figure 7) and v + , as well as the maximum value of v sense , must be considered to ensure that the senselo pin doesnt exceed the range listed in the electrical characteristics table. the senselo and sensehi pins of the lt6109 can function from 0.2v figure 6. gain error vs resistor tolerance sensehi lt6109 i sense r sense v + 9 v ? 5 v + r in v batt senselo 10 1 outa 8 610912 f05 r out v out r in + ? + figure 5. r in + reduces error due to i b output voltage error, ?v out(gain error) , due to external resistors the lt6109 exhibits a very low gain error. as a result, the gain error is only signifcant when low tolerance resistors are used to set the gain. note the gain error is systematically negative. for instance, if 0.1% resistors are used for r in and r out then the resulting worst-case gain error is C0.4% with r in = 100. figure 6 is a graph of the maximum gain error which can be expected versus the external resistor tolerance. output current limitations due to power dissipation the lt6109 can deliver a continuous current of 1ma to the outa pin. this current fows through r in and enters the current sense amplifer via the sensehi pin. the power dissipated in the lt6109 due to the output signal is: p out = (v sensehi C v outa ) ? i outa since v sensehi v + , p outa (v + C v outa ) ? i outa resistor tolerance (%) 0.01 0.01 resulting gain error (%) 0.1 1 10 0.1 1 10 610912 f06 r in = 100 r in = 1k
LT6109-1/lt6109-2 15 610912fa above the positive supply to 33v below it. these operat - ing voltages are limited by internal diode clamps shown in figures 1 and 2. on supplies less than 35.5v, the lower range is limited by v C + 2.5v. this allows the monitored supply, v batt , to be separate from the lt6109 positive supply as shown in figure 7. figure 8 shows the range of operating voltages for the senselo and sensehi inputs, for different supply voltage inputs (v + ). the senselo and sensehi range has been designed to allow the lt6109 to monitor its own supply current (in addition to the load), as long as v sense is less than 200mv. this is shown in figure 9. minimum output voltage the output of the lt6109 current sense amplifer can produce a non-zero output voltage when the sense voltage is zero. this is a result of the sense amplifer v os being forced across r in as discussed in the output voltage er - ror, ?v out(vos) section. figure 10 shows the effect of the input offset voltage on the transfer function for parts at the v os limits. with a negative offset voltage, zero input sense voltage produces an output voltage. with a positive offset voltage, the output voltage is zero until the input sense voltage exceeds the input offset voltage. neglect - ing v os , the output circuit is not limited by saturation of pull-down circuitry and can reach 0v. response time the lt6109 amplifer is designed to exhibit fast response to inputs for the purpose of circuit protection or current monitoring. this response time will be affected by the external components in two ways, delay and speed. applications information figure 9. lt6109 supply current monitored with load figure 7. v + powered separately from load supply (v b att ) figure 8. allowable senselo, sensehi voltage range sensehi lt6109 i sense r sense v + 9 v ? 5 v + r in v batt senselo 10 1 outa 8 610912 f07 r out v out ? + 60 50 40 30 20 20.2v 40.2v 10 27 allowable operating voltages on senselo and sensehi inputs (v) 2.8v 2.5v 2.7 10 20 30 35.5 40 50 v + (v) 60 610912 f08 valid senselo/ sensehi range sensehi lt6109 i sense r sense v + 9 v ? 5 r in v batt senselo 10 1 outa 8 610912 f09 r out v out ? + figure 10. amplifer output voltage vs input sense voltage input sense voltage (v) 0 output voltage (mv) 40 80 120 20 60 100 200 400 600 800 610912 f10 1000 1000 300 500 700 900 v os = ?125v v os = 125v g = 100
LT6109-1/lt6109-2 16 610912fa if the output current is very low and an input transient occurs, there may be an increased delay before the output voltage begins to change. the typical performance characteristics show that this delay is short and it can be improved by increasing the minimum output current, either by increasing r sense or decreasing r in . note that the typical performance characteristics are labeled with respect to the initial sense voltage. the speed is also affected by the external components. using a larger r out will decrease the response time, since v out = i outa ? z out where z out is the parallel combination of r out and any parasitic and/or load capacitance. note that reducing r in or increasing r out will both have the effect of increasing the voltage gain of the circuit. if the output capacitance is limiting the speed of the system, r in and r out can be decreased together in order to maintain the desired gain and provide more current to charge the output capacitance. the response time of the comparators is the sum of the propagation delay and the fall time. the propagation delay is a function of the overdrive voltage on the input of the comparators. a larger overdrive will result in a lower propagation delay. this helps achieve a fast system response time to fault events. the fall time is affected by the load on the output of the comparator as well as the pull-up voltage. the lt6109 amplifer has a typical response time of 500ns and the comparators have a typical response time of 500ns. when confgured as a system, the amplifer output drives the comparator input causing a total system response time which is typically greater than that implied by the individually specifed response times. this is due to the overdrive on the comparator input being determined by the speed of the amplifer output. internal reference and comparators the integrated precision reference and comparators com- bined with the high precision current sense allow for rapid and easy detection of abnormal load currents. this is often critical in systems that require high levels of safety and reliability. the lt6109 comparators are optimized for fault detection and are designed with latching outputs. latch- ing outputs prevent faults from clearing themselves and applications information require a separate system or user to reset the outputs. in applications where the comparator output can intervene and disconnect loads from the supply, latched outputs are required to avoid oscillation. latching outputs are also useful for detecting problems that are intermittent. the comparator outputs on the lt6109 are always latching and there is no way to disable this feature. each of the comparators has one input available externally, with the two versions of the part differing by the polarity of those available inputs. the other comparator inputs are connected internally to the 400mv precision reference. the input threshold (the voltage which causes the output to transition from high to low) is designed to be equal to that of the reference. the reference voltage is established with respect to the device v C connection. comparator inputs the comparator inputs can swing from v C to 60v regardless of the supply voltage used. the input current for inputs well above the threshold is just a few pas. with decreas - ing input voltage, a small bias current begins to be drawn out of the input near the threshold, reaching 50na max when at ground potential. note that this change in input bias current can cause a small nonlinearity in the outa transfer function if the comparator inputs are coupled to the amplifer output with a voltage divider. for example, if the maximum comparator input current is 50na, and the resistance seen looking out of the comparator input is 1k, then a change in output voltage of 50v will be seen on the analog output when the comparator input voltage passes through its threshold. if both comparator inputs are con - nected to the output then they must both be considered. setting comparator thresholds the comparators have an internal precision 400mv refer - ence. in order to set the trip points of the LT6109-1 com - parators, the output currents, i over and i under , as well as the maximum output current, i max , must be calculated: i over = v sense(over) r in , i under = v sense(under) r in , i max = v sense(max) r in
LT6109-1/lt6109-2 17 610912fa where i over and i under are the over and under currents through the sense resistor which cause the comparators to trip. i max is the maximum current through the sense resistor. depending on the desired maximum amplifer output volt - age (v max ) the three output resistors, r1, r2 and r3, can be confgured in two ways. if: v max > 400mv i over + 400mv ?i under r1 ( ) i under ? ? ? ? ? ? i max then use the confguration shown in figure 3. the desired trip points and full-scale analog output voltage for the circuit in figure 3 can then be achieved using the follow - ing equations: applications information r1 = 400mv i over r2 = 400mv ? i under r1 ( ) i under r3 = v max ? i max r1 + r2 ( ) i max if: v max < 400mv i over + 400mv ?i under r1 ( ) i under ? ? ? ? ? ? i max then use the confguration shown in figure 11. figure 11. typical confguration with alternative r out confguration outa i outa ? + ? + v + c1 sensehi inc2 inc1 r1 610912 f11 v ? v + v + v ? LT6109-1 senselo en/rst outc2 v reset r c v pullup load v supply v sense r sense undercurrent flag overcurrent flag r in + ? outc14 3 2 1 5 ? + v ? v ? v + i sense = v sense r sense r c r2 6 7 8 9 10 r3 c l v out 400mv reference c lc c lc
LT6109-1/lt6109-2 18 610912fa applications information the desired trip points and full-scale analog output voltage for the circuit in figure 13 can be achieved as follows: r1 = 400mv i over r2 = v max ? i max r1 ( ) i max r3 = 400mv ? i under r1 + r2 ( ) i under trip points for the lt6109-2 can be set by replacing i under with a second overcurrent, i over2 . hysteresis each comparator has a typical built-in hysteresis of 10mv to simplify design, ensure stable operation in the pres - ence of noise at the inputs, and to reject supply noise that might be induced by state change load transients. the hysteresis is designed such that the threshold voltage is altered when the output is transitioning from low to high as is shown in figure 12. external positive feedback circuitry can be employed to increase the effective hysteresis if desired, but such circuitry will have an effect on both the rising and fall - ing input thresholds, v th (the actual internal threshold remains unaffected). figure 13 shows how to add additional hysteresis to a noninverting comparator. r6 can be calculated from the extra hysteresis being added, v hys(extra) and the amplifer output current which you want to cause the comparator output to trip, i under . note that the hysteresis being added, v hys(extra) , is in addition to the typical 10mv of built-in hysteresis. r6 = 400mv ? v hys(extra) i under r1 should be chosen such that r1 >> r6 so that v outa does not change signifcantly when the comparator trips. figure 12. comparator output transfer characteristics figure 13. noninverting comparator with added hysteresis v hys outc1 (LT6109-1/lt6109-2) outc2 (lt6109-2) outc2 (LT6109-1) v hys v th increasing v inc1,2 610912 f12 ? + v + v + v ? inc2 v ? 5 610912 f13 outa 7 8 v + v + sensehi 9 10 1 3 LT6109-1 r in r sense i load r3 v + senselo outc2 400mv reference r5 r6 r1 vth r2 + ?
LT6109-1/lt6109-2 19 610912fa r3 should be chosen to allow suffcient v ol and compara - tor output rise time due to capacitive loading. r2 can be calculated: r2 = r1 ? v + ? 400mv ( ) ? v hys(extra) ? r3 ( ) v hys(extra) for very large values of r2 pcb related leakage may become an issue. a tee network can be implemented to reduce the required resistor values. the approximate total hysteresis will be: v hys = 10mv + r1 ? v + ? 400mv r2 + r3 ? ? ? ? ? ? for example, to achieve i under = 100a with 50mv of total hysteresis, r6 = 3.57k. choosing r1 = 35.7k, r3 = 10k and v + = 5v results in r2 = 4.12m. the analog output voltage will also be affected when the comparator trips due to the current injected into r6 by the positive feedback. because of this, it is desirable to have (r1 + r2 + r3) >> r6. the maximum v outa error caused by this can be calculated as: ? v outa = v + ? r6 r1 + r2 + r3 + r6 ? ? ? ? ? ? applications information in the previous example, this is an error of 4.3mv at the output of the amplifer or 43v at the input of the amplifer assuming a gain of 100. when using the comparators with their inputs decoupled from the output of the amplifer, they may be driven directly by a voltage source. it is useful to know the threshold voltage equations with the additional hysteresis. the input falling edge threshold which causes the output to transition from high to low is: v th(f) = 400mv ? r1 ? 1 r1 + 1 r2 + r3 ? ? ? ? ? ? ? v + ? r1 r2 + r3 ? ? ? ? ? ? the input rising edge threshold which causes the output to transition from low to high is: v th(r) = 410mv ? r1 ? 1 r1 + 1 r2 ? ? ? ? ? ? figure 14 shows how to add additional hysteresis to an inverting comparator. r7 can be calculated from the amplifer output current which is required to cause the comparator output to trip, i over . r7 = 400mv i over , assuming r1+r2 ( ) >> r7 figure 14. inverting comparator with added hysteresis ? + v + v + v ? inc1 v ? 5 610912 f14 outa 8 9 6 v + v + sensehi LT6109-1 r in r sense i load v + senselo outc14 1 10 400mv reference r3 r6 r7 r1 vth r2 v dd ? +
LT6109-1/lt6109-2 20 610912fa applications information to ensure (r1 + r2) >> r7, r1 should be chosen such that r1 >> r7 so that v outa does not change signifcantly when the comparator trips. r3 should be chosen to allow suffcient v ol and compara - tor output rise time due to capacitive loading. r2 can be calculated: r2 = r1 ? v dd ? 390mv v hys(extra) ? ? ? ? ? ? note that the hysteresis being added, v hys(extra) , is in addition to the typical 10mv of built-in hysteresis. for very large values of r2 pcb related leakage may become an issue. a tee network can be implemented to reduce the required resistor values. the approximate total hysteresis is: v hys = 10mv + r1 ? v dd ? 390mv r2 ? ? ? ? ? ? for example, to achieve i over = 900a with 50mv of total hysteresis, r7 = 442. choosing r1 = 4.42k, r3 = 10k and v dd = 5v results in r2 = 513k. the analog output voltage will also be affected when the comparator trips due to the current injected into r7 by the positive feedback. because of this, it is desirable to have (r1 + r2) >> r7. the maximum v outa error caused by this can be calculated as: ? v outa = v dd ? r7 r1 + r2 + r7 ? ? ? ? ? ? in the previous example, this is an error of 4.3mv at the output of the amplifer or 43v at the input of the amplifer assuming a gain of 100. when using the comparators with their inputs decoupled from the output of the amplifer they may be driven directly by a voltage source. it is useful to know the threshold voltage equations with additional hysteresis. the input rising edge threshold which causes the output to transi- tion from high to low is: v th(r) = 400mv ? 1 + r1 r2 ? ? ? ? ? ? the input falling edge threshold which causes the output to transition from low to high is: v th(f) = 390mv ? 1 + r1 r2 ? ? ? ? ? ? ? v dd r1 r2 ? ? ? ? ? ? comparator outputs the comparator outputs can maintain a logic low level of 150mv while sinking 500a. the outputs can sink higher currents at elevated v ol levels as shown in the typical performance characteristics. load currents are conducted to the v C pin. the output off-state voltage may range between 0v and 60v with respect to v C , regardless of the supply voltage used. as with any open-drain device, the outputs may be tied together to implement wire-or logic functions. the LT6109-1 can be used as a single-output window comparator in this way. en/rst pin the en/rst pin performs the two functions of resetting the latch on the comparators as well as shutting down the lt6109. after powering on the lt6109, the comparators must be reset in order to guarantee a valid state at their outputs. applying a pulse to the en/ rst pin will reset the compara- tors from their tripped state as long as the input on the comparator is below the threshold and hysteresis for an inverting comparator or above the threshold and hysteresis for a noninverting comparator. for example, if v inc1 is pulled higher than 400mv and latches the comparator, a reset pulse will not reset that comparator unless its input is held below the threshold by a voltage greater than the 10mv typical hysteresis. the comparator outputs typically unlatch in 0.5s with 2pf of capacitive load. increased capacitive loading will cause increased unlatch time. figure 15 shows the reset functionality of the en/ rst pin. the width of the pulse applied to reset the compara - tors must be greater than t rpw(min) (2s) but less than t rpw(max) (15s). applying a pulse that is longer than 40s typically (or tying the pin low) will cause the part to enter shutdown. once the part has entered shutdown, the supply current will be reduced to 3a typically and the amplifer, comparators and reference will cease to function
LT6109-1/lt6109-2 21 610912fa applications information until the en/ rst pin is transitioned high. when the part is disabled, both the amplifer and comparator outputs are high impedance. when the en/ rst pin is transitioned from low to high to enable the part, the amplifer output pmos can turn on momentarily causing typically 1ma of current to fow into the sensehi pin and out of the outa pin. once the amplifer is fully on, the output will go to the correct cur - rent. figure?16 shows this behavior and the impact it has on v outa . circuitry connected to outa can be protected from these transients by using an external diode to clamp v outa or a capacitor to flter v outa . power up after powering on the lt6109, the comparators must be reset in order to guarantee a valid state at their outputs. fast supply ramps may cause a supply current transient during start-up as shown in the typical performance characteristics. this current can be lowered by reducing the edge speed of the supply. reverse-supply protection the lt6109 is not protected internally from external rever - sal of supply polarity. to prevent damage that may occur during this condition, a schottky diode should be added in series with v (figure 17). this will limit the reverse current through the lt6109. note that this diode will limit the low voltage operation of the lt6109 by effectively reducing the supply voltage to the part by v d . also note that the comparator reference, comparator output and en/rst input are referenced to the v C pin. in order to preserve the precision of the reference and to avoid driving the comparator inputs below v C , r2 must connect to the v C pin. this will shift the amplifer output voltage up by v d . v outa can be accurately measured differentially across r1 and r2. the comparator output low voltage will also be shifted up by v d . the en/rst pin threshold is referenced to the v C pin. in order to provide valid input levels to the lt6109 and avoid driving en/ rst below v C the negative supply of the driving circuit should be tied to v C . figure 16. amplifer enable response figure 15. comparator reset functionality en/rst outc1 outc2 t reset 0.5s (typical) 610912 f15 t rpw(max) 15s comparator reset reset pulse width limits t rpw(min) 2s 50s/div 0v v en/rst 2v/div 0v v outa 2v/div 610912 f14 v + = 60v r in = 100 r out = 10k
LT6109-1/lt6109-2 22 610912fa applications information applications information ? + v + v + v ? inc v ? 5 v d + ? v outa + ? 610912 f17 outa 8 9 6 v + v + sensehi LT6109-1 r in r sense i load v dd v dd senselo outc4 en/rst 2 1 10 400mv reference r3 r1 r2 v dd ? + figure 17. schottky prevents damage during supply reversal typical applications overcurrent and undervoltage battery fault protection sensehi senselo outa 0.1 r10 100 LT6109-1 v ? to load v out 9.53k 475 30v undervoltage detection 0.8a overcurrent detection inc1 inc2 v + en/rst outc2 8 1 6 7 5 outc1 10 9 10k 100k 6.2v* irf9640 2 5v inc2 3 4 reset 6109 ta02 100k 2n7000 *cmh25234b 1m 13.3k + + + + 0.1f 10f 12 lithium 40v cell stack the comparators monitor for overcurrent and undervolt - age conditions. if either fault condition is detected the battery will immediately be disconnected from the load. the latching comparator outputs ensure the battery stays disconnected from the load until an outside source resets the lt6109 comparator outputs.
LT6109-1/lt6109-2 23 610912fa typical applications mcu interfacing with hardware interupts sensehi senselo outa 0.1 v + 100 LT6109-1 v ? to load v out adc in 2k 6.65k inc2 1.33k inc1 v + en/rst outc1 8 1 7 6 5 outc2 10 9 10k 2 5v 4 3 reset 6109 ta03 10k 5v v out /adc in atmega1280 pb0 pb1 pcint2 pcint3 adc2 pb5 5 6 7 2 3 1 undercurrent routine reset comparators mcu interupt outc2 goes low 5v 0v 610912 ta03b example: the comparators are set to have a 50ma undercurrent threshold and a 300ma overcurrent threshold. the mcu will receive the comparator outputs as hardware interrupts and immediately run an appropriate fault routine. simplifed dc motor torque control the fgure shows a simplifed dc motor control circuit. the circuit controls motor current, which is proportional to motor torque; the lt6109 is used to provide current feedback to a difference amplifer that controls the current in the motor. the ltc ? 6992 is used to convert the output of the difference amp to the motors pwm control signal. sensehi senselo outa lt6109 v ? 5.62k 100k 2 3 7 ltc6246 1f 4 6 1 3 6 irf640 5v 1n5818 0.1 v motor 5 4 2 0.47f v out current set point (0v to 5v) 3.4k 1k inc2 1m 610912 ta04 1k 78.7k 100k 280k inc1 v + en/rst outc1 outc2 reset 100f + ? ltc6992-1 v + gnd 5v brushed dc motor (0a to 5a) mabuchi rs-540sh mod set out div
LT6109-1/lt6109-2 24 610912fa typical applications typical applications power-on reset or disconnect using a timerblox ? circuit ? + ? + v ? v + v + v ? ? + v ? 5 inc1 610912 ta06 v + 400mv reference v + r in 100 r sense i load r5 10k inc2 outa 6 7 8 9 sensehi LT6109-1 5v senselo outc2 outc1 en/rst 10 1 3 4 2 r1 8.06k r2 1.5k r3 499 r4 10k r7 1m r6 487k r8 30k trig c1 0.1f q1 2n2222 out gnd v + set div ltc6993-3 5v creates a delayed 10s reset pulse on start-up optional: discharges c1 when supply is disconnected the ltc6993-1 provides a 10s reset pulse to the lt6109 - 1. the reset pulse is delayed by r7 and c1 whose time constant must be greater than 10ms and longer than the supply turn-on time. optional components r8 and q1 discharge capacitor c1 when the supply and/or ground are discon - nected. this ensures that when the power supply and/or ground are restored, capacitor c1 can fully recharge and trigger the ltc6993-3 to produce another comparator reset pulse. these optional components are particularly useful if the power and/or ground connections are intermittent, as can occur when pcb are plugged into a connector.
LT6109-1/lt6109-2 25 610912fa typical applications precision power-on reset using a timerblox ? circuit ? + ? + v ? v + v + v ? ? + v ? 5 inc1 610912 ta07 v + 400mv reference v + r in 100 r sense i load r5 10k inc2 outa 6 7 8 sensehi LT6109-1 9 5v senselo outc2 outc1 en/rst 10 1 3 4 2 r8 100k r1 8.06k r2 1.5k r3 499 r4 10k r4 487k c2 0.1f r5 681k r6 1m 10s reset pulse generator c1 0.1f r7 191k 1 second delay on start-up trig out gnd v + set div ltc6993-1 trig gnd set ltc6994-1 out v + div
LT6109-1/lt6109-2 26 610912fa package description ms package 10-lead plastic msop (reference ltc dwg # 05-08-1661 rev e) msop (ms) 0307 rev e 0.53 0.152 (.021 .006) seating plane 0.18 (.007) 1.10 (.043) max 0.17 ?0.27 (.007 ? .011) typ 0.86 (.034) ref 0.50 (.0197) bsc 1 2 3 4 5 4.90 0.152 (.193 .006) 0.497 0.076 (.0196 .003) ref 8910 7 6 3.00 0.102 (.118 .004) (note 3) 3.00 0.102 (.118 .004) (note 4) note: 1. dimensions in millimeter/(inch) 2. drawing not to scale 3. dimension does not include mold flash, protrusions or gate burrs. mold flash, protrusions or gate burrs shall not exceed 0.152mm (.006") per side 4. dimension does not include interlead flash or protrusions. interlead flash or protrusions shall not exceed 0.152mm (.006") per side 5. lead coplanarity (bottom of leads after forming) shall be 0.102mm (.004") max 0.254 (.010) 0 ? 6 typ detail ?a? detail ?a? gauge plane 5.23 (.206) min 3.20 ? 3.45 (.126 ? .136) 0.889 0.127 (.035 .005) recommended solder pad layout 0.305 0.038 (.0120 .0015) typ 0.50 (.0197) bsc 0.1016 0.0508 (.004 .002)
LT6109-1/lt6109-2 27 610912fa information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no representa - tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. revision history rev date description page number a 12/12 addition of a-grade performance and electrical characteristics correction to typical application diagram addition of a-grade order information clarifcation to absolute maximum short circuit duration edits to electrical characteristics conditions and notes clarifcation to nomenclature used in typical performance characteristics clarifcation to description of pin functions internal reference block redrawn for consistency edits to applications information addition of lt6108 to related parts 1, 3, 4, 11, 13, 15 (fig10), 28 1 2 2 3, 4 5-8 8, 9 9, 10, 12, 17, 18, 19, 25, 26 10-16, 18, 20-25 28
LT6109-1/lt6109-2 28 610912fa linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 fax : (408) 434-0507 www.linear.com linear technology corporation 2011 lt 1212 rev a ? printed in usa related parts typical application adc driving application part number description comments lt1787 bidirectional high side current sense amplifer 2.7v to 60v, 75v offset, 60a quiescent, 8v/v gain ltc4150 coulomb counter/battery gas gauge indicates charge quantity and polarity lt6100 gain-selectable high side current sense amplifer 4.1v to 48v, gain settings: 10, 12.5, 20, 25, 40, 50v/v ltc6101 high voltage high side current sense amplifer up to 100v, resistor set gain, 300v offset, sot-23 ltc6102 zero drift high side current sense amplifer up to 100v, resistor set gain, 10v offset, msop8/dfn ltc6103 dual high side current sense amplifer 4v to 60v, resistor set gain, 2 independent amps, msop8 ltc6104 bidirectional high side current sense amplifer 4v to 60v, separate gain control for each direction, msop8 lt6105 precision rail-to-rail input current sense amplifer C0.3v to 44v input range, 300v offset, 1% gain error lt6106 low cost high side current sense amplifer 2.7v to 36v, 250v offset, resistor set gain, sot-23 lt6107 high temperature high side current sense amplifer 2.7v to 36v, C55c to 150c, fully tested: C55c, 25c, 150c lt6108 high side current sense amplifer with reference and comparator 2.7v to 60v, 125v offset, resistor set gain, 1.25% threshold error lt6700 dual comparator with 400mv reference 1.4v to 18v, 6.5a supply current sensehi senselo outa 0.1 sense low sense high LT6109-1 v ? out 2k 0.1f 0.1f 6.65k inc2 1.33k overcurrent undercurrent inc1 v + en/rst outc1 8 1 7 6 5 outc2 10 9 2 4 v cc v cc v ref 10k 3 reset 6109 ta05 in v cc 10k in + ltc2470 comp to mcu 100 the low sampling current of the ltc2470 16-bit delta sigma adc is ideal for the lt6109.

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